The present application relates to subject matters described in co-pending application Ser. No. 09/672,688 filed on Sep. 29, 2000.
The present invention generally relates to a transmitter. More particularly, the present invention is concerned with a negative feedback amplifier circuit employed in the transmitter of digital modulation type for compensating for nonlinear distortions. Further, the invention is also concerned with a method of controlling the phase of the negative feedback amplifier circuit.
In radio systems in which a linear digital modulation system such as, for example, 16QAM (Quadrature Amplitude Modulation), xcfx80/4 shift QPSK (Quadrature Phase Shift Keying) or the like is employed, it is indispensably required to compensate for nonlinear distortion of a power amplifier. To this end, a variety of nonlinear distortion compensating systems (linearizers) are proposed for practical application. Among them, a Cartesian loop negative feedback type linearizer has been conventionally employed long since. For having better understanding of the background techniques of the present invention, description will first be made in some detail of the conventional linear feedback amplifier known heretofore by reference to FIG. 2 which is a block diagram showing an arrangement of a transmitting section of a digital radio system provided with a Cartesian loop negative feedback type linearizer.
Referring to FIG. 2, reference numeral 1 denotes a baseband signal generator which is designed to output an in-phase component (hereinafter referred to simply as the I-component) and a quadrature component (hereinafter referred to simply as the Q-component) of a baseband signal. The I-component is added with a corresponding feedback signal by an adder 2-1, the output of which is supplied to a loop filter 3-1. Likewise, the Q-component is added with a corresponding feedback signal by an adder 2-2, the output signal of which is applied to a loop filter 3-2. The loop filers 3-1 and 3-2 operate to limit the bandwidths of the inputted I-component and the inputted Q-component, respectively. The I- and Q-components undergone the bandwidth limitation are then inputted to a quadrature modulator 4, as indicated by Ixe2x80x2 and Qxe2x80x2, respectively.
A numeral 11 denotes a reference signal generator which is designed to generate a reference frequency signal which is then supplied to the first and second PLL frequency synthesizers 12 and 13, respectively. The PLL frequency synthesizer 12 is designed to generate a first local oscillation signal (hereinafter referred to as the first LO1 signal) on the basis of the reference frequency signal. The first LO1 signal is then supplied to a quadrature modulator 4 and a phase shifter 18. On the other hand, the PLL frequency synthesizer 13 generates a second local oscillation signal (hereinafter referred to as the second LO2 signal) on the basis of the reference signal. The second LO2 signal is supplied to mixers 6 and 15, respectively. The phase shifter 18 controls the phase of the first LO1 signal in conformance with a control signal supplied from a phase controller 19. The first LO1 signal undergone the phase control is then supplied to a quadrature demodulator 16.
The quadrature modulator 4 serves to orthogonally modulate the first LO1 signal (a carrier signal) into a signal of an intermediate frequency band (hereinafter referred to as the IF frequency band) with the I-component Ixe2x80x2 and the Q-component Qxe2x80x2 of the baseband signal inputted to the quadrature modulator 4. Then, the modulated signal is applied to a bandpass filter (BPF) 5. The bandpass filter 5 operates to eliminate unnecessary components from the modulated signal. The output signal of the bandpass filter 5 is then applied to the mixer 6. The mixer 6 operates to convert the modulated signal applied therein into a signal of a desired frequency by making use of the second LO2 signal outputted from the PLL frequency synthesizer 13. The output signal of the mixer 6 is then applied to a bandpass filter (BPF) 7. The bandpass filter 7 serves to eliminate unnecessary spurious components from the signal inputted. The output of the bandpass filter 7 is then inputted to the amplifying circuit (PA) 8 which operates to amplify the input signal to a specified or rated output level for transmission by way of an antenna 9.
Since the negative feedback amplifier described above is implemented in the form of the negative feedback linearizer based on the Cartesian loop, a part of the output signal of the amplifying circuit 8 is fed back to the input-side circuitry to be supplied to an attenuator (ATT) 14 through the medium of a directivity coupler 10. In response, the attenuator 14 operates to regulate the power level of the input signal to a proper value. The output of the attenuator 14 is supplied to the mixer 15. The mixer 15 then converts the frequency of the signal inputted from the attenuator 14 to an IF frequency by using the second LO2 signal. The IF frequency signal is then supplied to the quadrature demodulator 16.
The quadrature demodulator 16 operates to divide the inputted IF signal into two IF signals having 90xc2x0-phase shifted from each other and produce baseband signals i and q of the I-component and the Q-component, respectively, by making frequency conversion of two IF signals with the first LO1 signal inputted from the phase shifter 18 which has undergone the phase control. The I-component is applied to a subtracting or minus input terminal of the adder 2-1 as the I-component baseband signal i for feedback by way of a switch 20-1, while the Q-component is applied to a subtracting or minus input terminal of the adder 2-2 as the Q-component baseband signal q for feedback by way of a switch 20-2. In this manner, feedback operation is performed on the I-component and the Q-component, respectively. At this time, the input terminals a and b of the switches 20-1 and 20-2 are connected to the first and second adders 2-1 and 2-2, respectively.
In this negative feedback amplifier circuit, it is required for the purpose of circuit stabilization that the input signals I and Q on one hand and the feedback signals i and q on the other hand are in phase, respectively, (i.e., phase difference=0xc2x0) on the input side of the adders 2-1 and 2-2. To say in another way, in case phase difference takes place between the input signals and the feedback signals, it is required that such control be carried out that the phase of the first LO signal inputted to the quadrature demodulator 16 can be adjusted by 180xc2x0 (xcfx80 radian) at maximum by means of the phase shifter 18 in order to make the phases of the input signals coincide with those of the feedback signals in the adders 2-1 and 2-2, respectively.
At this juncture, a phase control method will be described below. Initially, the switches 20-1 and 20-2 shown in FIG. 2 are set to such positions that the outputs q and i of the quadrature demodulator 16 are supplied to the phase controller 19 with the feedback loop being in the opened state.
In this state, in the baseband signal generator 1, a predetermined DC voltage is given to only the I-component for the purpose of phase adjustment while the Q-component being held zero (Q=0), whereon the quadrature modulation is carried out straightforwardly for signal transmission by way of the antenna 9 in accordance with the procedure described previously. In that case, the output waveform of the amplifying circuit 8 assumes the waveform of the non-modulated carrier signal. Such being the circumstances, when a part of the output of the amplifying circuit 8 is fed back by way of the directivity coupler 10, then the DC voltage makes appearance only for the I-component of the feedback signal outputted from the quadrature demodulator 16 while no DC voltage makes appearance for the Q-component so long as the feedback signals i and q outputted from the quadrature demodulator 16 are in phase with each other. By contrast, when the output signals i and q of the quadrature demodulator 16 are out of phase with each other, a DC voltage corresponding to the phase deviation between these output signals appears on the side of the Q-component. Thus, the angle of rotation corresponding to the phase deviation can be determined on the basis of the DC voltages of the I-component and the Q-component.
In the phase controller 19, the phase corresponding to the angle of rotation as determined is rotated in the direction reverse to that of the phase deviation by controlling correspondingly the phase shifter 18 to thereby adjust the phase of the first LO1 signal so that the phases of the feedback signals outputted from the quadrature demodulator 16 are in phase with the phases of the input signals. In this manner, the negative feedback loop can be stabilized. When the phases of the input signals coincide with those of the feedback signals, the output of the Q-component becomes zero. Thus, the switches 20-1 and 20-2 can be changed over to the adders 2-1 and 2-2, respectively, at this time point. Now, the closed loop operation becomes effective.
As will now be understood from the foregoing, in the case of the conventional negative feedback power amplifier, the feedback loop has to be opened every time the phase adjusting operation is to be effectuated. However, so long as the transmitter is operating continuously, the feedback loop remains in the closed state. Consequently, the phase adjustment cannot be performed for the change of phase during the transmitting operation. Besides, the switches 20-1 and 20-2 are employed for opening/closing the feedback loop, and the phase is controlled on the basis of the DC voltage of the feedback signal on the input side of the these change-over switches. Consequently, it is required to perform offset adjustment on the input side of the change-over switches. In this conjunction, it is however noted that when the feedback loop is closed by means of the switches 20-1 and 20-2 after adjustment of the offset voltage in the open loop state, the offset voltage will become deviated from the adjusted level due to voltage drops brought about by turn-on resistances across the switches 20-1 and 20-2, respectively. In other words, the offset adjustment can not accurately be performed, which of course means that the phase control with sufficient accuracy is practically impossible.
If the transmitting operation is continued with the offset remaining as it is, then the DC offset which is one of the transmission performance factors will undergo deterioration. Furthermore, when the initial phase setting is performed with the offset remaining deviated, the transmitting operation is carried out with the phase being left mismatched. As a result of this, sufficient phase margin can no more be assured, presenting a cause for generation of spurious components, leading ultimately to deterioration of the transmission characteristics.
In Gailus et al""s, U.S. Pat. No. 5,066,923 issued on Nov. 19, 19991, there is disclosed a method of performing the phase adjustment by opening the feedback loop in such manner as described above by reference to FIG. 2. Furthermore, in the specification of Japanese Patent No. 2746133 assigned to NEC corporation, a technique for adjusting the phase in the closed loop state is disclosed. However, the technique taught by the patent mentioned just above, both the I-component and the Q-component are required indispensably as the feedback signals for effectuating the phase adjustment, wherein phase comparison of the input baseband I-component with the feedback I-component as well as the phase comparison of the input baseband Q-component with the feedback Q-component has to be carried out, which will incur much complication in the circuit configuration, needless to say.
In the light of the state of the art described above, it is an object of the present invention to provide a power amplifier circuit incorporating a negative feedback circuit for a transmitter, which is so arranged that the phase adjustment can be performed without need for opening the feedback loop with a simplified circuit configuration while protecting the transmission characteristics against deterioration and thus ensuring constantly stable output operation characteristics.
Another object of the present invention is provide a phase control method for the power amplifier circuit mentioned above.
In view of the above and other object which will become apparent as the description proceeds, there is provided according to an aspect of the present invention a power amplifier circuit and a phase control method for the same, the gist of which resides in that an I-component test signal for an I-component baseband signal and a Q-component test signal for a Q-component baseband signal are supplied as inputs to a feedback loop. The I-component test signal and an I-component baseband feedback signal are added together by an adder to generate an I-component summing signal while the Q-component test signal and a Q-component baseband feedback signal are added together by another adder for generating a Q-component summing signal. A quadrature modulator orthogonally modulates a carrier signal generated by an oscillator with the I-component summing signal and the Q-component summing signal. A power amplifier amplifies the orthogonally modulated signal. A quadrature demodulator orthogonally demodulates the orthogonally modulated signal by using a part of the amplified signal and the carrier signal to thereby generate an I-component baseband feedback signal and a Q-component baseband feedback signal. The quadrature modulator changes the phase of the carrier signal generated by the oscillator in accordance with a phase control signal. A phase controller compares either one of the I-component summing signal or the Q-component summing signal as selected with a reference signal to thereby generate a signal indicative of result of the comparison. Further the phase controller generates the phase control signal for changing sequentially and incrementally the phase of the carrier signal in a predetermined direction. The phase of the carrier signal is fixed at the time point when the signal indicative of result of the comparison meets a predetermined condition in the course of changing the phase of the carrier signal.
To say in another way, in the state where the feedback loop is operating with the test signals being inputted thereto, the phase of the carrier signal is changed gradually or incrementally while monitoring the signals inputted to the quadrature modulator. Then, the operating condition under which the phases of the input baseband signals coincide with those of the feedback signals can be detected without fail. The phase of the carrier signal is then fixed to the phase value corresponding to the time point at which the operating condition mentioned above is detected. Subsequently, the ordinary transmitting operation can be carried out with the fixed phase.
In a preferred mode for carrying out the phase control method according to the present invention, DC voltages differing each other may be used as the I-component test signal and the Q-component test signal, respectively. By way of example, the DC voltages which satisfy the conditions that I=1 and Q=0 in an I-Q orthogonal coordinate system can be employed as the I- and Q-baseband signal inputs, respectively. The voltage value of either the I-component signal or the Q-component signal inputted to the quadrature modulator is compared with a reference voltage value (which meets the conditions that I=0 and that Q=0) in the course of rotating the phase of the carrier signal clockwise or counterclockwise in the I-Q orthogonal coordinate system, to thereby acquire a polarity signal which can assume positive (plus) polarity or negative (minus) polarity. By detecting the time point at which the polarity signal changes from the plus to minus polarity or vice versa (i.e., the time point when the conditions that I=1 and that Q=0 are met), it is possible to detect the point at which the phases of the input I- and Q-component signals coincide with those of the feedback signals, respectively.
The above and other objects, features and attendant advantages of the present invention will more easily be understood by reading the following description of the preferred embodiments thereof taken, only by way of example, in conjunction with the accompanying drawings.